RF Power detector

ABSTRACT

A method and apparatus is provided for detecting the output power of an RF power amplifier for purposes of controlling the output power. A circuit for generating an output power control signal includes a power detector to detect the output power of an RF power amplifier. A variable gain amplifier is coupled to the power detector for amplifying the output of the power detector. The value of the generated control signal is a function of the gain of the variable gain amplifier.

FIELD OF THE INVENTION

[0001] This invention relates to the field of power amplifiers. Moreparticularly, this invention relates to circuitry for detecting theoutput power of an RF power amplifier.

BACKGROUND OF THE INVENTION

[0002] In some applications utilizing a power amplifier, it is desirableto limit the peak voltage that the switching devices of the poweramplifier are subjected to. For example, in CMOS devices, the transistorbreakdown voltage may be only slightly greater than the supply voltage.Therefore, CMOS devices are not well suited to traditional poweramplifier designs, where switching devices are subjected to voltages atleast twice the supply voltage.

[0003]FIG. 1 is a schematic diagram of a conventional Class E amplifier.As shown, a transistor M1 is connected between ground and an inductor L1which is connected to a voltage source V_(dd). The gate of thetransistor M1 is connected to an input signal Vi. The connection of thetransistor M1 and the inductor L1 forms a node labeled Vd. The switchingdevice M1, as well as other switching devices described may be comprisedof any suitable switching devices, for example, MOSFETs or othertransistor types. A capacitor C1 is connected between Vd and ground. Theamplifier includes a transformation network consisting of inductor L2and capacitor C2. The capacitor C2 is connected to a load R_(L) atoutput node V_(o).

[0004]FIG. 2 is a timing diagram illustrating the input signal Vi andthe resulting voltage at Vd. As shown, the input signal Vi is a squarewave signal switching between ground and V_(dd). When the input signalVi is high (V_(dd)), the transistor M1 is turned on, holding Vd toground. When the input signal Vi transitions to low, transistor M1 turnsoff and the voltage at Vd rises above V_(dd). During this time, thetransistor M1 must sustain this high drain-to-source voltage. Afterpeaking, the voltage at Vd decreases until it reaches ground. In atypical prior art Class E design, this peak voltage is approximately 3.6V_(dd). Although the peak voltage can be reduced slightly, it can not bedecreased below about 2.5 V_(dd) since the average voltage at Vd mustequal V_(dd). Designs such as that shown in FIG. 1 are not well suitedto certain device technologies, such as CMOS, where transistor breakdownvoltages are only slightly higher than the supply voltage.

[0005] It can therefore be seen that there is a need for amplifierdesigns where the peak voltages applied to the transistors of theamplifier are reduced so that they are below the transistor breakdownvoltages of the devices being used to implement the design.

[0006] Another problem relating to amplifiers relates to the use ofdifferential circuits. It is difficult to performdifferential-to-single-ended conversion when a single ended load isrequired with high efficiency. Therefore, there is a need for improveddifferential-to-single-ended conversion designs.

[0007] Another problem relating to amplifiers relates to detecting theoutput power of an amplifier for purposes of controlling the outputpower of the amplifier. For example, in a power regulation circuit for acellular telephone power amplifier, there is a need to sense the powerdelivered to the antenna. The sensed power is used to help control theoutput power of the power amplifier.

SUMMARY OF THE INVENTION

[0008] An apparatus of the invention is provided for a circuit forgenerating a control signal for controlling the output power of a poweramplifier of a wireless device by comprising: a power detector coupledto the output of the power amplifier for generating a detector outputsignal; and a variable gain amplifier coupled to the power detector foramplifying the detector output signal to a desired level, wherein thevalue of the control signal generated by the circuit is a function ofthe gain of the variable gain amplifier.

[0009] Another embodiment of the invention provides a method ofcontrolling the output power of an RF power amplifier comprising thesteps of: detecting the output power of the RF power amplifier andgenerating a first signal from the detected output power; amplifying thefirst signal using a variable gain amplifier; generating a gain controlsignal from the output of the amplifier and from a reference signal;using the gain control signal to set the gain of the amplifier;generating a second signal by conditioning the gain control signal; andusing the second signal to control the output power of the RF poweramplifier.

[0010] One embodiment includes an integrated circuit for use with anexternal directional coupler comprising: an RF power amplifier formed onthe integrated circuit, wherein the integrated circuit is configuredsuch that an external directional coupler can be used to generate adetector signal based on the output power of the RF power amplifier; anda power detector formed on the same integrated circuit to generate anoutput signal based on the detector signal.

[0011] Other objects, features, and advantages of the present inventionwill be apparent from the accompanying drawings and from the detaileddescription that follows below.

BRIEF DESCRIPTION OF THE DRAWINGS

[0012] The present invention is illustrated by way of example and notlimitation in the figures of the accompanying drawings, in which likereferences indicate similar elements and in which:

[0013]FIG. 1 is a schematic diagram of a prior art Class E amplifier.

[0014]FIG. 2 is a timing diagram illustrating the voltage at V_(D)relative to the input signal V_(I) for the prior art Class E amplifiershown in FIG. 1.

[0015]FIG. 3 is a block diagram illustrating an example of anenvironment in which a power amplifier of the present invention may beused.

[0016]FIG. 4 is a schematic diagram of one embodiment of a poweramplifier of the present invention.

[0017]FIG. 5 is a timing diagram illustrating the voltages present inthe amplifier shown in FIG. 4, relative to the input signals.

[0018]FIG. 6 is a schematic diagram of an embodiment of a poweramplifier of the present invention with a load connected differentially.

[0019]FIG. 7 is a schematic diagram of an embodiment of a poweramplifier of the present invention connected to a single-ended load.

[0020]FIG. 8 is a schematic diagram of an embodiment of a poweramplifier of the present invention connected differentially.

[0021]FIG. 9 is a timing diagram illustrating the voltages present inthe amplifier shown in FIG. 8.

[0022]FIG. 10 is a schematic diagram of an embodiment of a poweramplifier of the present invention.

[0023]FIG. 11 is a schematic diagram of another embodiment of a poweramplifier of the present invention.

[0024]FIG. 12 is a schematic diagram of an embodiment of a poweramplifier of the present invention having a preamplifier circuit.

[0025]FIG. 13 is a timing diagram illustrating the voltages present inthe amplifier shown in FIG. 12.

[0026]FIG. 14 is a schematic diagram of an embodiment of a two-stagedifferential power amplifier of the present invention.

[0027]FIG. 15 is a schematic diagram of a prior art circuit used forperforming differential-to-single-ended conversion.

[0028]FIG. 16 is a block diagram of a differential-to-single-endedconversion and impedance transformation circuit of the presentinvention.

[0029]FIG. 17 is a schematic diagram of a differential-to-single-endedconversion and impedance transformation circuit of the presentinvention.

[0030]FIGS. 18 and 19 are schematic diagrams illustrating differentialinputs AC-coupled from a load.

[0031]FIG. 20 is a block diagram of a differential-to-single-endedconversion and impedance transformation circuit having multipledifferential inputs.

[0032]FIG. 21 is a block diagram of a voltage regulator of the presentinvention.

[0033]FIG. 22 is a schematic diagram of an embodiment of a voltageregulator of the present invention.

[0034]FIG. 23 is a schematic diagram of an embodiment of a voltageregulator of the present invention.

[0035]FIG. 24 is a schematic diagram of an embodiment of a voltageregulator of the present invention.

[0036]FIG. 25 is an isometric view illustrating how a device of thepresent invention is packaged.

[0037]FIG. 26 is a side view of the device shown in FIG. 25.

[0038]FIG. 27 is a diagram illustrating a ceramic chip carrier with aninductor formed in the carrier.

[0039]FIG. 28 is a diagram illustrating a ceramic chip carrier with avertically-formed inductor formed in the carrier.

[0040]FIG. 29 is an electrical schematic diagram of inductors connectedbetween four connection points.

[0041]FIG. 30 is a diagram illustrating an example of how the inductorsshown in FIG. 29 could be formed in a ceramic chip carrier.

[0042]FIG. 31 is a schematic diagram of a prior art power detector.

[0043]FIG. 32 is a plot illustrating V_(SENSE) as a function of theoutput power of the amplifier shown in FIG. 31.

[0044]FIG. 33 is a block diagram of a circuit for controlling the outputpower of an RF power amplifier.

[0045]FIGS. 34 and 35 illustrate examples of circuitry for sensing andcontrolling the output power of an RF power amplifier.

[0046]FIG. 36 is a plot illustrating V_(SENSE) as a function of theoutput power of the amplifier shown in FIG. 33.

[0047]FIG. 37 is a plot illustrating the sense error for the circuitryshown in FIG. 34 at various output levels.

DETAILED DESCRIPTION

[0048] In order to provide a context for understanding this description,the following illustrates a typical application of the presentinvention. A power amplifier of the present invention may be used as anamplifier for use with a wireless transmission system such as a wirelesstelephone or other device. The invention may also be applied to otherapplications, including, but not limited to, RF power amplifiers. In awireless device such as a cellular telephone, the device may include atransceiver, an antenna duplexer, and an antenna. Connected between thetransceiver and the antenna duplexer is an RF power amplifier foramplifying signals for transmission via the antenna. This is one exampleof an application of a power amplifier of the present invention. Ofcourse the invention may be used in any other application requiring apower amplifier. In the case of a wireless telephone application, theinvention may be applied to GSM or other constant envelope modulationsystems.

[0049]FIG. 3 is a block diagram illustrating an example of anenvironment in which a power amplifier of the present invention may beused. FIG. 3 shows a power amplifier 310 connected to a pair of inputsignals V_(in) and V_(ip). The input signals come from an input 312 froman input network such as the transceiver chip mentioned above. An inputbuffer is formed by a plurality of inverters X1 and X2 which areconnected to the input 312 as shown. The input buffer circuit could alsobe comprised of more or less inverters, or any other suitable circuitry.The power amplifier 310 is also connected to a voltage regulator 314which provides a regulated voltage source V_(dd) from a voltage source,such as battery voltage VB. The power amplifier 310 is also connected toa transformation network 316 which is connected to a load 318. Note thatthe connection between power amplifier 310 and the transformationnetwork 316 may be comprised of a single or multiple connections. FIG. 3is shown with n connections. In the example of a wireless transmissionsystem, the load 318 may be comprised of an antenna. Note that thecomponents shown in FIG. 3 are optional and are not essential to thepower amplifier 310.

[0050]FIG. 4 is a schematic diagram of one embodiment of a poweramplifier of the present invention. The power amplifier includes aswitching device M1 connected between ground and the node labeledV_(dn). The gate of the switching device M1 is connected to the inputsignal V_(in). Another switching device M2 is connected between thevoltage source V_(dd) and a node labeled V_(dp). The gate of theswitching device M2 is connected to the input signal V_(ip). Connectedbetween the switching devices M2 and M1 is an inductor L1. FIG. 4 alsoshows a capacitor C1 connected between V_(dn) and ground. A capacitor C3is connected between V_(dp) and Vdd. The capacitors C1 and C3 may becomprised of a combination of separate capacitors and parasiticcapacitances of the switching devices M1 and M2. The power amplifiershown in FIG. 4 also includes a reactive network connected betweenV_(dn) and the amplifier output V_(o). The reactive network is formed byinductor L2 and capacitor C2 and can be used for filtering or impedancetransformation. A load R_(L) is connected to the amplifier output V_(o).

[0051] The power amplifier shown in FIG. 4 resembles a push-pullamplifier topologically, but is fundamentally different, in that theinput signals V_(in) and V_(ip) are inverses of one another. Sinceswitching device M1 is an n-channel device and switching device M2 is ap-channel device, the switching devices M1 and M2 are both turned on andturned off during the same time intervals. FIG. 5 is a timing diagramillustrating the voltages present in the amplifier shown in FIG. 4,relative to the input signals. FIG. 5 shows the input signals V_(in) andV_(ip) which are 180° out of phase with each other. In other words, whenone of the input signals is high, the other is low. During phase 1(V_(in) high and V_(ip) low), the switching devices M1 and M2 are bothturned on so that V_(dp) and V_(dn) are clamped to V_(dd) and groundrespectively. During phase 2 (V_(in) low and V_(ip) high), the switchingdevices M1 and M2 are both turned off. The voltage at V_(dn) rises andbegins to ring at a frequency determined by the values of the componentsL1, C1, C3, L2, and C2. For the best efficiency, these components arechosen so that V_(dn) rises and then returns to ground immediatelybefore the end of phase 2. The voltage at V_(dp) falls and rings in asimilar way. The voltage at node V_(dp) rises back to V_(dd) immediatelybefore the end of phase 2, when switching devices M1 and M2 are turnedon.

[0052] The peak voltages present across the switching devices M1 and M2can be adjusted as desired by changing the passive component values inthe circuit under the constraint that the average voltage of V_(dn) mustequal that of V_(dp). If this average voltage lies at V_(dd)/2 then thepeak value of V_(dn) will be only slightly higher than V_(dd) and thatof V_(dp) will be only slightly lower than ground. The duty cycle of theinput signals V_(in) and V_(ip) waveforms can be adjusted to reduce thepeak voltages even further. As a result, this configuration eliminatesthe large signal swings that transistors are subjected to in the priorart.

[0053] The power amplifier shown in FIG. 4 does not take full advantageof the signal swing that occurs on node V_(dp). An increase inefficiency can be achieved by making use of the signal swing on bothV_(dp) and V_(dn). This can be accomplished by connecting the loaddifferentially across nodes V_(dp) and V_(dn) as shown in FIG. 6. FIG. 6shows a power amplifier similar to that shown in FIG. 4. The poweramplifier includes switching devices M1 and M2, inductor L1, andcapacitors C1 and C3. A transformation network 616 is connected to bothnodes V_(dp) and V_(dn). A load R_(L) is connected to the transformationnetwork 616. The waveforms for the power amplifier shown in FIG. 6 aresimilar to those for the power amplifier shown in FIG. 4. In thisembodiment, the current flowing through the load R_(L) is determined bythe difference between the voltages on V_(dp) and V_(dn).

[0054] When a single-ended load is required, the transformation networkcan be made to facilitate a single-ended load. FIG. 7 shows a poweramplifier with two capacitors C2 and C4 and an inductor L3 connected asshown between V_(dn) and V_(o). An inductor L2 is connected betweenV_(dp) and the connection point of the capacitors C2 and C4. Asingle-ended load R_(L) is connected between V_(o) and ground. Thewaveforms for the power amplifier shown in FIG. 7 are similar to thosefor the power amplifier shown in FIG. 4. In this embodiment, the currentflowing to the output from V_(dp) and current flowing to the output fromV_(dn) add when they are summed in phase at the load. The load is ACcoupled from either V_(dp) or V_(dn) by capacitor C4. The inductor L2and capacitor C2 can also be chosen to transform the load impedanceR_(L) into a desired impedance so that power delivered to the load canbe adjusted independently from the voltage swing on Vdp and Vdn. In thiscase, the voltage swing on V_(o) will vary from that on V_(dp) andV_(dn) as determined by the selection of C2 and L2. Since thecombination of L2 and C2 is a tuned circuit, it provides some bandpassfiltering. If additional filtering is desired, capacitor C4 and inductorL3 can also be used as an additional bandpass filter. In summary, L2 andC2 in the configuration of FIG. 7 simultaneously perform the functionsof impedance transformation, filtering, and differential-to-single-endedconversion.

[0055] The amplifier of the present invention may also be implementeddifferentially using two amplifiers (such as the amplifier shown in FIG.7) connected together as shown in FIG. 8. FIG. 8 shows a first amplifier(the positive side) comprised of switching devices M1+ and M2+, inductorL1+, capacitors C1+ and C3+, and a transformation network comprised ofcapacitors C2+ and C4+ and inductors L2+ and L3. A second amplifier (thenegative side) is comprised of switching devices M1− and M2−, inductorL1−, capacitors C1− and C3−, and a transformation network comprised ofcapacitors C2− and C4− and inductors L2− and L3. The two amplifiers aresimilar to each other with the inductors L2 and capacitors C2interchanged as shown. The input signals V_(in−) and V_(ip−) on thenegative side are shifted by 180 degrees from the input signals V_(in+)and V_(ip+) on the positive side. FIG. 9 is a timing diagramillustrating the voltages present at the nodes V_(dn+), V_(dp+),V_(dn−), and V_(dp−).

[0056] The values of the passive components in the amplifier shown inFIG. 8 may be chosen so that the resulting currents from both amplifierssum in phase at the load R_(L). The advantages of the power amplifiershown in FIG. 8 are similar to the advantages common to differentialcircuits in general. For example, undesired interference from supply orsubstrate noise is common-mode. Another advantage is that the impact ofsupply resistance is reduced because the supply current flows duringboth clock phases.

[0057] Note that the load R_(L) shown in FIG. 8 could be connected toonly two of the four output nodes of the power amplifier. For example, aconfiguration similar to that shown in FIG. 4 could be connecteddifferentially to the load R_(L), where the nodes V_(dp+) and V_(dp−)are not connected to V_(o).

[0058]FIG. 8 also shows an alternate embodiment where an optionalinductor L4 is connected (shown in dashed lines) between nodes V_(dp+)and V_(dp−). Without the optional inductor L4, the voltage swings onnodes Vdp+, Vdp−, Vdn+ and Vdn− and the values of capacitors C1+, C1−,C3+ and C3− can not be independently adjusted. The optional inductor L4has the advantage that these voltage swings can be adjustedindependently of the capacitance values mentioned above.

[0059] The capacitors C1 and C3 described above are used to shape thewaveforms of the voltages on V_(dp) and V_(dn). As mentioned above,these capacitances may be provided by separate capacitors or by theparasitic capacitances of switching devices M1 and M2. In anotherembodiment, these capacitances are formed by switching devices in a waythat improves the efficiency of the amplifier.

[0060]FIG. 10 is a schematic diagram of a power amplifier similar to theamplifier shown in FIG. 8. In the amplifier shown in FIG. 10, thecapacitors C1+ and C3+ are replaced by switching devices M3− and M4−,respectively. Similarly, the capacitors C1− and C3− are replaced byswitching devices M3+ and M4+, respectively. Each of the switchingdevices M3 and M4 are driven as shown by a voltage from the oppositeamplifier. For example, the switching device M4+ is driven by thevoltage at V_(dp−) on the negative side. The switching device M4− isdriven by the voltage at V_(dp+) on the positive side. Similarly, theswitching device M3+ is driven by the voltage at V_(dn−) while theswitching device M3− is driven by the voltage at V_(dn+). The waveformsfor the amplifier shown in FIG. 10 are similar to those described above.

[0061] The amplifier shown in FIG. 10 allows the switching devices M1+and M1− to be made smaller by an amount equal to the size of switchingdevices M3+ and M3−. Similarly, the switching devices M2+ and M2− can bemade smaller by an amount equal to the size of switching devices M4+ andM4−. However, switching devices M1 and M2 should remain sufficientlylarge to assure stability of the circuit. A decrease in the size of theswitching devices M1 and M2 improves the efficiency since the inputcapacitances that must be driven are smaller. Another advantage to theamplifier shown in FIG. 10 is that cross-coupling helps to assure thatthe waveforms present at V_(dp−) and V_(dn−) have the correct phaserelationship to the waveforms present at V_(dp+) and V_(dn+), despitepossible timing variations on the positive inputs (V_(ip+), V_(in+)) andon the negative inputs (V_(ip−), V_(in−)).

[0062]FIG. 10 also shows an alternate embodiment where an optionalinductor L4 is connected (shown in dashed lines) between nodes V_(dp+)and V_(dp−), similar to the inductor L4 shown in FIG. 8. If the optionalinductor L4 is connected, the voltage swings of nodes Vdp+, Vdp−, Vdn+,and Vdn− can be chosen independently from the input capacitances of M4−,M4+, M3−, M3+.

[0063]FIG. 11 is a schematic diagram of a power amplifier similar to theamplifier shown in FIG. 10, but with the inductors L1+ and L1− replaced.Inductor L1+ is replaced with a pair of inductors L1A+ and L1B+.Inductor L1− is replaced with a pair of inductors L1A− and L1B−. Thenode formed by the connection of inductors L1A+ and L1B+ is connected tothe node formed by the connection of inductors L1A− and L1B−. Theembodiment shown in FIG. 11 has similar advantages to the embodiment inFIG. 10 with the optional inductor L4 in that it allows the voltageswings of nodes Vdp+, Vdp−, Vdn+, and Vdn− to be chosen independentlyfrom the input capacitances of M4−, M4+, M3−, M3+.

[0064] As described above with respect to FIG. 3, input buffer circuitrymay be used to drive the gates of the switching devices M1 and M2 of theamplifiers described above. However, the efficiency may be improved if asimilar amplifier circuit is used as a preamplifier circuit. FIG. 12 isan example of an amplifier having a preamplifier circuit.

[0065]FIG. 12 shows an amplifier similar to the amplifier shown in FIG.7. At the input of the amplifier, a preamplifier is shown. Thepreamplifier is comprised of switching devices M5 and M6 connectedbetween ground and V_(dd). An inductor L3 is connected between theswitching devices M5 and M6. The preamplifier includes inputs V_(ip2)and V_(in2). The preamplifier circuit receives input signals V_(ip2) andV_(in2) and generates signals V_(ip) and V_(in) for use by theamplifier. The preamplifier circuit is similar to the amplifiersdescribed above, except that all of the passive elements except inductorL3 are eliminated. The capacitances required by the preamplifiercircuitry are formed from the input capacitances of the gates ofswitching devices M1 and M2. Of course, other passive elements could beused with the preamplifier circuit.

[0066]FIG. 13 is a timing diagram illustrating the waveforms at V_(in),V_(ip), V_(dn), and V_(dp) of FIG. 12. The preamplifier output waveformsV_(ip) and V_(in) have a shape that makes them well suited for drivingthe input gates of switching devices M1 and M2 in the final stage.

[0067] Note that in an alternate configuration the capacitor C4 could beconnected between inductor L2 and V_(o) with capacitor C2 connectedbetween V_(dn) and V_(o). This alternate configuration functionssimilarly to the configuration shown in FIG. 12.

[0068]FIG. 14 is a schematic diagram of an amplifier using a two-stagedifferential configuration which provides an increased efficiency overthe circuit shown in FIG. 12. The amplifier shown in FIG. 14 is similarto the differential amplifier shown in FIG. 10, with the addition ofpreamplifier circuitry. The inputs V_(ip+) and V_(in+) of the amplifierare connected to preamplifier circuitry comprised of switching devicesM5+ and M6+. The switching devices M5+ and M6+ are connected betweenground and V_(dd), with an inductor L3+ connected between them.Capacitances are provided to nodes V_(dp2+) and V_(dn2+) by switchingdevices M8+ and M7+, respectively. The negative side of the amplifier isconfigured in the same manner. The positive and negative sides of thepreamplifier circuitry are cross-coupled in the same way as theamplifier circuitry shown in FIG. 10 (described above). In thisconfiguration, the input capacitances of the NMOS and PMOS switchingdevices M1 and M2 of the power amplifier, the input capacitances of thepreamplifier switching devices M7 and M8, and the value of inductor L5can be adjusted so that the signals at V_(dp2) and V_(dn2) have thedesired peak amplitudes.

[0069] Another aspect of the present invention relates to a circuit andmethod of providing differential-to-single ended output conversion andimpedance transformation from differential signals. Differentialcircuits have a number of advantages that are well known. For example,the impact from noise sources is reduced since these signals arecommon-mode (i.e., the positive and negative sides are effectedidentically). In addition, even-order harmonics are reduced because ofcircuit symmetry. Because of these and other advantages, a differentialconfiguration may be desirable even when the load is single-ended. If asingle-ended load is needed, circuitry for differential-to-single-endedconversion is needed.

[0070] One prior art method for performing differential-to-single-endedconversion at high frequency involves use of a transformer or balun.FIG. 15 shows a prior art circuit used for performingdifferential-to-single-ended conversion using a transformer T1. Theprimary side of the transformer T1 is connected to a first differentialinput V₊ and a second differential input V⁻. The secondary side of thetransformer T1 is connected to ground and an output node V_(O). A loadZ_(L) is connected between ground and the output node V_(O). If thetransformer has a 1-to-1 turns ratio, then the differential signals V₊and V⁻ are translated into a signal having an amplitude of (V₊−V⁻)across the load Z_(L).

[0071] In some applications, impedance matching or impedancetransformation is needed to transform a given load impedance into adifferent impedance seen by the driver. Impedance transformation can beaccomplished, as part of the differential-to-single ended conversion,using the transformer circuit shown in FIG. 15 by adjusting the windingratio of the transformer T1. However, the use of transformers fordifferential-to-single-ended conversion and impedance transformation hasdisadvantages. First, high quality transformers are larger and morecostly than other passive elements and are not easily integrated withother semiconductor circuits. Second, practical transformers haveimperfect magnetic coupling which causes a loss of power from input tooutput.

[0072] The present invention provides a technique that performsdifferential-to-single ended conversion as well as impedancetransformation and avoids the disadvantages of a transformer solution.FIG. 16 shows a block diagram of a differential-to-single-endedconversion and impedance transformation circuit. The circuit has a firstimpedance X₁ coupled between the second differential input signal V⁻ andan output node V_(O). A second impedance X₂ is coupled between the firstdifferential input signal V₊ and the output node V_(O). A load Z_(L) isconnected between the output node V_(O) and ground. In the circuit shownin FIG. 16, current flowing to the output node V_(O) from differentialinput V₊ is shifted in phase from the voltage on V₊. Similarly, currentflowing to the output node V_(O) from differential input V⁻ is shiftedin phase from the voltage on V⁻. The impedances X1 and X2 are chosen sothat these two currents add together when they are summed at the loadZ_(L). For example, if X1 shifts the output current by +90 degrees andX2 shifts the output current by −90 degrees then the resultant currentswill sum in phase at the load. FIG. 17 illustrates one example of animplementation of the circuit shown in FIG. 16. FIG. 17 shows an L-Cdifferential-to-single-ended conversion and impedance transformationcircuit. The impedance X1 is comprised of a capacitor C5 which iscoupled between the second differential input signal V⁻ and the outputnode V_(O). The impedance X2 is comprised of an inductor L6 which iscoupled between the first differential input signal V₊ and the outputnode V_(O).

[0073] Referring back to FIG. 16, since the inputs V₊ and V⁻ aredifferential, the inputs have opposite signs. However, the differentialinputs V₊ and V⁻ are not necessarily equal in amplitude. The outputvoltage V_(O) of the differential-to-single-ended conversion andimpedance transformation circuit is given by the following equation:$\begin{matrix}{V_{O} = {\frac{\left( {{V_{+}X_{1}} + {V_{-}X_{2}}} \right)\left( {{{- j}\quad X_{2}X_{1}} + {\left( {X_{1} + X_{2}} \right)Z_{L}}} \right)}{\left( {\left( {X_{1}X_{2}} \right)^{2} + {\left( {X_{1} + X_{2}} \right)^{2}Z_{L}^{2}}} \right)}{Z_{L}.}}} & (1)\end{matrix}$

[0074] The power P_(L) delivered to the load Z_(L) is given by thefollowing equation: $\begin{matrix}{P_{L} = {\frac{\left( {{V_{+}X_{1}} + {V_{-}X_{2}}} \right)^{2}}{\left( {\left( {X_{1}X_{2}} \right)^{2} + {\left( {X_{1} + X_{2}} \right)^{2}Z_{L}^{2}}} \right)}{Z_{L}.}}} & (2)\end{matrix}$

[0075] Differential-to-single-ended conversion is achieved if theimpedances X₁ and X₂ have opposite signs. Impedances X₁ and X₂ may becomprised of any combination of reactive elements (e.g., capacitor C5and inductor L6 shown in FIG. 17) whose combination meets thisrequirement. For example, if differential inputs V₊ and V⁻ have equalamplitudes A, and impedances X₁ and X₂ have equal amplitudes X, then theoutput voltage V_(O) can be determined by substituting these values intoequation (1) above. The resulting output voltage V_(O) is given by thefollowing equation: $\begin{matrix}{V_{O} = {{- {j2A}}{\frac{Z_{L}}{X}.}}} & (3)\end{matrix}$

[0076] It can be seen from equation (3) that the ratio R/X can be chosenso that the amplitude of the output V_(O) is either larger or smallerthan the amplitude A of the differential input. The voltage of theoutput V_(O) increases as the value of X decreases. Similarly, thevoltage of the output V_(O) decreases as the value of X increases.

[0077] In certain applications, the load Z_(L) must be AC-coupled fromone of the differential inputs V⁻ or V₊. FIGS. 18 and 19 show examplesof a how the differential inputs may be AC-coupled from the load Z_(L)in the example shown in FIG. 17. In the circuit shown in FIG. 18, anadditional capacitor C6 is inserted between the output node V_(O) andboth the capacitor C5 and the inductor L6. The capacitor C6 AC-couplesthe output node V_(O) from the first and second differential inputs V₊and V⁻. In the circuit shown in FIG. 19, an additional capacitor C6 isinserted between the output node V_(O) and the inductor L6. Thecapacitor C6 AC-couples the output node V_(O) from the firstdifferential input V₊. Note that the capacitor C1 provides AC-couplingbetween the output node V_(O) from the second differential input V⁻.

[0078] The techniques for providing differential-to-single-endedconversion and impedance transformation described above can be appliedto circuits having multiple differential inputs. FIG. 20 shows adifferential-to-single-ended conversion and impedance transformationcircuit having multiple differential inputs. FIG. 20 shows differentialinputs V₁ through V_(N), where N is the total number of differentialinputs. A first impedance X₁ is coupled between the differential inputV₁ and the output node V_(O). A second impedance X₂ is coupled betweenthe differential input V₁ and the output node V_(O). Similarly, an Nthimpedance X_(N) is coupled between the differential input V_(N) and theoutput node V_(O). Each of the currents from each differential input issummed in phase at the output node V_(O). In this embodiment, theimpedance X_(j) between the jth differential input V_(j) and the outputnode V_(O) will depend on its phase with respect to that of otherdifferential inputs. Optimal power transfer to the load Z_(l) occurswhen the impedances X_(j) are purely reactive. However, this techniquemay still be applied when impedance X_(j) is not purely reactive. Forexample, this might occur when actual inductors and capacitors have aseries resistance.

[0079] As mentioned above, the RF power amplifier shown in FIG. 3includes a voltage regulator 314 connected between the power amplifier310 and a source of battery voltage VB to provide a voltage source VDD.In one embodiment of the present invention, the voltage regulator 314resides on the same integrated circuit as the power amplifier circuit.The function of the voltage regulator is to provide a source of voltageto the power amplifier and to help control the output power level. Forexample, in a cellular phone environment, a base station may dictate thepower level at which each cell phone should transmit (based on factorssuch as the physical distance from the base station, for example).Varying the voltage level (VDD) can control the output power of thepower amplifier. As the voltage of the voltage source VDD increases, theoutput power increases. Therefore, by controlling the operation of thevoltage regulator, and therefore controlling the voltage of voltagesource VDD, the output power of the amplifier can be controlled. Whilethe power amplifier 310 will function with any suitable voltageregulator or voltage source, described below is a detailed descriptionof a suitable voltage regulator.

[0080]FIG. 21 is a block diagram of a voltage regulator 544 used toprovide a regulated voltage VDD from a voltage source VB, for example,from a battery. As shown, the regulated voltage VDD is provided to adevice 530. The device 530 may be any type of device requiring a voltagesource including, but not limited to power amplifiers. The voltageregulator 544 includes an input 546 that is connected to a controlsignal VSET to control the voltage level VDD provided to the device 530.Following is a detailed description of the voltage regulator of thepresent invention in the context of its use in an RF power amplifier(such as that shown in FIG. 3). However, it is understood that thevoltage regulator may be used with any type of amplifier as well as anyother type of device requiring a voltage source.

[0081]FIG. 22 is a schematic diagram of a first embodiment of a voltageregulator 644 connected to a battery voltage VB. The voltage regulator644 is comprised of a device M9 and an op amp X4. The op amp X4 includesa first input 646 for connection to a voltage control signal VSET. In apreferred embodiment, the control signal VSET is an analog voltagesignal that is proportional to the desired voltage level. The otherinput to the op amp X4 is connected to the regulated voltage VDD. Theoutput of the op amp X4 is connected to the input of the device M9.

[0082]FIG. 23 is a schematic diagram of another embodiment of a voltageregulator 744 connected to a battery voltage VB. The voltage regulator744 is similar to the voltage regulator 644 shown in FIG. 22 with theaddition of a second regulator circuit comprised of op amp X5, switchingdevice M10, and an external resistor R1. FIG. 23 also shows anintegrated circuit 770 (dashed lines) to illustrate that the poweramplifier is formed on the integrated circuit 770 while the resistor R1is not. The integrated circuit 770 may also be the same integratedcircuit on which the device to be powered resides.

[0083] The first regulator circuit is connected in the same manner asthe regulator circuit shown in FIG. 22. The op amp X5 of the secondregulator circuit includes an input VSET2 for connection to a voltagecontrol signal. The other input to the op amp X5 is connected to theregulated voltage VDD. The output of the op amp X5 is connected to thegate of the device M10. The external resistor R1 is connected betweenthe battery voltage VB and the device M10. FIG. 23 also shows voltagecontrol circuitry 776 which has an input 746 connected to the controlsignal VSET. The voltage control circuitry 776 uses the signal VSET tocreate voltage control signals VSET1 and VSET2 for use by the first andsecond regulator circuits. By controlling both regulators, the voltagelevel VDD can be controlled. In addition, by selectively activating thesecond regulator, power can be dissipated off the integrated circuit 770(via resistor R1). This results in a reduction of heat generated in theintegrated circuit 770.

[0084] The voltage regulator 744 operates as follows. Since it isdesired to minimize the amount of power dissipated on the integratedcircuit 770, one goal is to maximize the use of the second regulatorcircuit (X5, M10) in order to maximize power dissipation through theexternal resistor R1. Therefore, voltage control circuitry 776 willenable the second regulator circuit to provide as much power as it canbefore enabling the first regulator circuit (X4, M9). In other words,when more power is required than the second regulator circuit canprovide, the first regulator circuit is enabled to provide additionalpower. In this way, the maximum amount of power will be dissipatedthrough external resistor R1.

[0085]FIG. 24 is a schematic diagram of another embodiment of voltageregulator 844 having multiple regulator s and multiple externalresistors. The voltage regulator 844 is similar to the regulator 744shown in FIG. 23, with the addition of a third regulator circuitcomprised of device M11, op amp X6, and external resistor R2. The thirdregulator circuit is connected in the same ways as the second regulatorcircuit, and operates in a similar manner. The op amp X6 of the thirdregulator circuit includes an input VSET3 for connection to a voltagecontrol signal. The other input to the op amp X5 is connected to theregulated voltage VDD. The output of the op amp X6 is connected to thegate of device M11. The external resistor R2 is connected between thebatter voltage VB and device M11. FIG. 24 also shows voltage controlcircuitry 876 which has an input 846 connected to the control signalVSET. The voltage control circuitry 876 uses the signal VSET to createvoltage control signals VSET1, VSET2, and VSET3 for use by the regulatorcircuits. By activating the second or third regulator, power can bedissipated off the integrated circuit 870 (via resistor R1 and/or R2).This results in a reduction of heat generated in the integrated circuit870.

[0086] The voltage regulator 844 operates as follows. Since it isdesired to minimize the amount of power dissipated on the integratedcircuit 870, one goal is to maximize the use of the second and thirdregulator circuits in order to maximize power dissipation through theexternal resistors R1 and R2. Therefore, voltage control circuitry 876will enable the second and third regulator circuits to provide as muchpower as it can before enabling the first regulator circuit. In otherwords, when more power is required than the second and/or thirdregulator circuit can provide, the first regulator circuit is enabled toprovide additional power. In this way, the maximum amount of power willbe dissipated through external resistors R1 and R2.

[0087] The values of the resistors R1 and R2 may be equal, or may bedifferent, depending on the needs of a user. In addition, the inventionis not limited to the use of one or two external resistors. Additionalregulator circuits and external resistors could be added. In oneembodiment, the value of resistor R1 is 0.7 ohms and the value ofresistor R2 is 0.3 ohms.

[0088] Another benefit of the present invention involves the use of dualgate oxide devices. In CMOS digital systems, it is sometimes desired toprovide devices suitable for use with two voltage levels (e.g., 3.3volts and 5 volts). Therefore, processing technologies have beendeveloped to provide a single integrated circuit having both 0.5 μm and0.35 μm devices. As mentioned above, a thicker gate oxide results in adevice with a higher breakdown voltage. On the other hand, a thinnergate oxide results in a faster device, but with a lower breakdownvoltage.

[0089] The RF amplifier of the present invention takes advantage of theavailability of dual gate oxide devices by selectively choosing certaingate lengths for various components of the amplifier. For example, ithas been discovered that for preprocessing circuitry or pre-drivercircuitry, a high speed is desirable and breakdown voltage is not asimportant. Therefore these devices are designed using a thinner gateoxide. For output state devices, where a high breakdown voltage is moreimportant, the devices are designed using a thicker gate oxide.

[0090] In one embodiment, a dual gate oxide device is used to create anRF amplifier such as the RF amplifier shown in FIGS. 12, and 14. Onesuitable use of dual gate oxides in these amplifiers is to utilizedevices having channel lengths of both 0.5 μm and 0.35 μm. The 0.5 μmand 0.35 μm devices have gate oxide thicknesses of 140 Angstroms (Å) and70 Å, respectively. Referring to the example shown in FIG. 12, thepredriver devices M5 and M6 can be chosen with much smaller devicewidths than the output devices M1 and M2. In this case, the predriveroutput signals Vip and Vin are nearly sinusoidal, the voltage difference(Vip−Vin) varies between about +Vdd and −Vdd, and the input capacitancesof M1 and M2 can be chosen so that neither M5 nor M6 experiences avoltage drop that is larger than Vdd. As a result, a high breakdownvoltage is not critical for the predriver and devices M5 and M6 can beimplemented using 0.35 μm devices. When high efficiency is desired,switching devices M1 and M2 of the final amplifier stage are sized withlarge device widths so that nodes Vdn and Vdp are strongly clamped totheir respective supply voltages of ground and Vdd when these devicesare on. In this case, the voltage difference (Vdp−Vdn) varies over arange that is larger than that of the predriver and either M1, M2, orboth will experience a voltage drop that is larger than Vdd. Since ahigher breakdown voltage is desired from these devices, M1 and M2 caneach be implemented using 0.5 μm devices. Since PMOS transistors aretypically slower than NMOS transistors and thicker gate oxide devicesare slower than thinner gate oxide devices, it is preferable to use athicker gate oxide for NMOS devices than for PMOS devices. An example ofthe use of dual gate oxide thicknesses for the RF amplifier of FIG. 14includes only NMOS devices with a thick gate oxide. Predrivertransistors M5+, M5−, M6+, M6−, M7+, M7−, M8+, and M8− are implementedusing 0.35 μm devices because, as described above, they are notsubjected to voltage drops greater than Vdd and breakdown is not acritical concern. As described above, the final amplifier stageexperiences larger voltage swings. However these larger swings can bedistributed across its NMOS and PMOS devices in such a way that onlyNMOS devices see a voltage swing larger than Vdd. This is accomplishedby adjusting the values of inductors L1+, L1−, and L4 and the inputcapacitances of devices M3+, M3−, M4+, and M4−. In this approach, PMOSdevices M2+, M2−, M4+, and M4− in the final amplifier stage are thinnergate oxide devices, whereas NMOS devices M1+, M1−, M3+, M3− are thickergate oxide devices.

[0091] Of course, the present invention is not limited to the valuesdescribed above. For example, as thinner gate oxides become more common,one or both thicknesses may become lower. In addition, note that theterms “thicker” or “thinner” in this description are intended to onlyrefer to intentional or significant differences in gate oxidethicknesses. For example, the 0.35 μm devices may vary from one anotherby some small amount depending on manufacturing tolerances. A 0.5 μmdevice is considered to be “thicker” than a 0.35 μm device. Also notethat this invention applies to various CMOS devices and that the RFAmplifier described above is only used as one example of the applicationof dual gate oxide devices of the present invention.

[0092] Another benefit of the present invention relates to how an RFpower amplifier of the present invention is packaged. The design of anRF amplifier requires a low inductance and low resistance to thetransistors or switching devices. In addition, RF power amplifierdesigns typically require a number of passive components such asinductors and capacitors. It is advantageous to integrate thesecomponents in the power amplifier package. The packaging technique ofthe present invention addresses these concerns by using “flip chip”technology and multi-layer ceramic chip carrier technology.

[0093]FIGS. 25 and 26 are isometric and side views, respectively,illustrating a packaging technique of the present invention. FIGS. 25and 26 show a “flip chip” integrated circuit 970 mounted to amulti-layer ceramic chip carrier 972. The integrated circuit 970includes a plurality of connection points, or “bumps” 974 on theunderside of the integrated circuit 970. Similarly, the ceramic chipcarrier 972 includes a plurality of connection points or bumps 976. Thebumps 974 of the integrated circuit 970 are formed by solder and can bemounted to corresponding conductive material formed on the upper surfaceof the ceramic chip carrier 972. Similarly, the bumps 976 of the ceramicchip carrier 972 are also formed by solder and are used to mount thechip carrier 972 to a printed circuit board (not shown). A typical flipchip allows 250 μm spaced bumps. A typical chip carrier also allows 250μm spaced vias for connection to the flip chip bumps 974. In oneexample, 6×6 mm ceramic chip carrier includes 36 bumps 976 forconnection to a PCB. Flip chip and ceramic chip carrier technologies areconsidered conventional and will not be described in detail.

[0094] Various benefits can be realized by selectively placing certaincomponents of the RF power amplifier of the present invention onintegrated circuit 970 and ceramic chip carrier 972. The invention willbe described with respect to the RF power amplifier shown in FIG. 14,although the invention is not limited to power amplifiers. In oneembodiment of the invention, all of the switching devices are formed onthe integrated circuit 970. In addition, the power transistors (such asswitching devices M1+, M1−, M2+, M2−) formed on the integrated circuit970 are preferably placed directly below the bumps 974 of the integratedcircuit 970 resulting in low resistance and inductance (as compared towire bond integrated circuit packages).

[0095] The multi-layer ceramic chip carrier 972 is used to build high-Qinductors, transformers, and capacitors. This can be beneficial for CMOSpower amplifier architecture since multiple inductors and capacitors maybe required. For example, a single band power amplifier may require 4-8inductors which would be impractical to build on a printed circuitboard. In addition, multiple matching networks are used to provide thehigh transformation ratio required in a push-pull amplifier design. Inone embodiment of the invention, the transformers, inductors,capacitors, and other passive devices are formed on the ceramic chipcarrier 972. The ceramic chip carrier 972 includes multiple conductivelayers 978 (shown as hidden lines) that can be designed to implementthese passive devices.

[0096] In one embodiment of the RF power amplifier shown in FIG. 14, allof the switching devices and capacitors C2+ and C2 reside on theintegrated circuit 970, with the inductors L3+, L3−, L5, L1+, L1−, L4,L2+, and L2− residing on the multi-layer ceramic chip carrier 972.

[0097] In a CMOS power amplifier design, multiple high-Q inductors arerequired to tune out large on-chip gate capacitances. Since thesecapacitances are large, the required inductors are low in value anddifficult to integrate. One solution is to form high-Q inductors on theceramic chip carrier. FIG. 27 is a diagram illustrating the ceramic chipcarrier 972 shown in FIGS. 25 and 26 with a horizontally-formed inductor1180 formed in the ceramic chip carrier 972. The inductor 1180 includesa horizontal loop portion formed by conductive tracel 182 connected totwo bumps 974 of the ceramic chip carrier 972 by two vias 1184. Onedisadvantage with the inductor 1180 is that the inductor connectionpoints needs to be close to the edge of the ceramic chip carrier 972unless the value of the inductor is large enough to route to a lowerlayer of the ceramic chip carrier 972.

[0098]FIG. 28 is a diagram illustrating the ceramic chip carrier 972with a vertically-formed inductor 1280 formed in the carrier 972. Theinductor 1280 is formed in the vertical direction by vias 1284 extendingto conductive trace 1286, which may be formed on a lower level of thecarrier 972. As shown, the inductor 1280 extends downward into theceramic chip carrier 972 and is coplanar, since the vias 1284 and trace1286 exist on the same plane. The vias 1284 may be formed throughseveral layers of the carrier 972, depending the inductance desired. Avertically-formed inductor such as the inductor 1280 has two majoradvantages over horizontally-formed inductors. First, thevertically-formed inductors can be formed underneath the chip 970without blocking other routing channels. Therefore, more layout optionsare available, and more inductors can be formed. Second, thevertically-formed vias 1284, as opposed to the horizontal conductivetrace 1182, result in less loss at RF frequencies since the vias 1284have a greater cross-sectional surface area than the conductive traces.The vias 1284 are substantially cylindrical and have a surface area ofπdL, where d is the diameter of the via 1284 (e.g., 100 μm) and L is thelength of the via. The conductive traces, such as conductive trace 1182,have a surface area of 2 dL. Therefore, the resistance of a via at RFfrequencies is approximately π/2 less than the resistance of aconductive trace 1182.

[0099]FIGS. 29 and 30 illustrate one embodiment of vertically-formedinductors of the present invention. FIG. 29 is an electrical schematicdiagram showing inductors L7, L8, L9, L10, and L11 connected betweenconnection points 1310, 1312, 1314, and 1316. As shown, inductors L7 andL8 are connected between connection points 1310 and 1312. Similarly,inductors L9 and L10 are connected between connection points 1314 and1316. Inductor L11 is connected between connection points 1318 and 1320,which are formed between inductors L9 and L10, and L7 and L8.

[0100]FIG. 30 illustrates an example of how the circuit of FIG. 29 canbe implemented using vertically-formed inductors of the presentinvention. The connection points 1310, 1312, 1314, and 1316 are formedat the surface of the ceramic chip carrier (not shown in FIG. 30) andwill be electrically connected to four of the bumps 974 of the flip-chip970. In this example, the inductors are formed using the upper twolayers of the ceramic chip carrier. Vias 1322 and 1324 extend throughboth layers where they are connected to an end of conductive traces 1326and 1328, respectively, formed in the lower layer of the ceramic chipcarrier. The opposite ends of the conductive traces 1326 and 1328 areconnected to vias 1330 and 1332, respectively, which are also formed inthe lower layer of the ceramic chip carrier. Together, the via 1322,conductive trace 1326, and via 1330 form inductor L7. Similarly, the via1324, conductive trace 1328, and via 1332 form inductor L9. The vias1330 and 1332 are connected to opposite ends of conductive trace 1334,formed in the upper layer. The conductive trace 1334 forms the inductorL11. Finally, vias 1336 and 1338 are connected to the vias 1330 and1332, respectively, as well as to opposite ends of the conductive trace1334. The vias 1336 and 1338 form the inductors L8 and L10,respectively. While FIGS. 29 and 30 show one specific example of howinductors could be formed in the ceramic chip carrier, it should beunderstood that other implementations are possible.

[0101] Another benefit of the present invention relates to sensing theoutput power of an RF power amplifier for purposes of controlling theoutput power. As mentioned above, in some devices (e.g., cellulartelephones or other wireless communication devices), there is a need tosense the power delivered to the antenna of a device so that the outputpower of the device can be precisely controlled.

[0102]FIG. 31 illustrates a prior art approach for detecting the outputpower of a power amplifier. FIG. 31 shows a circuit 3100 including apower amplifier 3110 and an antenna 3112 coupled to the output of thepower amplifier 3110. A directional coupler 3114 is coupled to sense theoutput power of the power amplifier 3110. The directional coupler 3114generates a signal V_(COUP) that is rectified by a Schotkey diode D1 andthen filtered by the RC filter (comprised of resistor R3 and capacitorC7). The rectified and filtered signal is provided as an input to alinear amplifier 3116. The amplifier 3116 generates a DC signalV_(SENSE) which may be used to control the output power of the poweramplifier 3110. The level of V_(SENSE) provides an indication of theamount of power provided to the antenna 3112. Generally, as the outputpower of the power amplifier 3110 increases, the voltage of V_(SENSE)also increases.

[0103]FIG. 32 is a plot of a curve illustrating V_(SENSE) as a functionof the output power (P_(ANTENNA)) of the power amplifier 3110. As shown,the curve is not linear, which can cause problems. The non-linearV_(SENSE) curve requires that each device produced (e.g., each cellphone) be calibrated at various power levels. This takes time andincreases the ultimate cost of the device. Another problem with thisprior art approach is that the circuitry is not accurate, especially atlower power levels. In addition, the temperature sensitivity of theSchotkey diode D1 effects the accuracy of the circuitry.

[0104] The present invention provides a solution to the problems foundin the prior art by approximating a linear V_(SENSE) curve on a powerlog scale. FIG. 33 is a block diagram of a circuit for controlling theoutput power of an RF power amplifier. FIG. 33 shows a power amplifier3310 and an antenna 3312 coupled to the output of the power amplifier3310. A power detector (shown in FIG. 33 as directional coupler 3314) iscoupled to the output of the power amplifier 3310 for sensing the outputpower of the power amplifier 3310. The directional coupler 3314generates a detector output signal V_(COUP) which is proportional to theoutput power of the power amplifier 3310. The signal V_(COUP) isprovided to a coupler variable gain amplifier (coupler VGA) 3318. Someexamples of suitable coupler VGAs are described below. The amplifiedoutput of the coupler VGA 3318 is provided to the input of a sensecircuit 3320. The sense circuit 3320 may be provided by a circuitry thatresponds to the envelope the carrier signal. Examples of circuitrysuitable for use as sense circuits include, but are not limited to, peakdetectors, RMS detector, rectifiers, etc. The output of the sensecircuit 3320 is provided to a first input of an op amp 3322. A secondinput of the op amp 3322 is coupled to a reference voltage (V_(SET)).Note that “V_(SET)” referred to with respect to FIGS. 33-35 is adifferent signal than “VSET” referred to with respect to the earlierfigures. The op amp 3322 generates a gain control signal which is fedback to the coupler VGA 3318 for controlling the gain of the coupler VGA3318. The gain control signal is also provided to a conditioning circuit3324 which conditions the gain control signal and generates a DC signalV_(SENSE). As the output power sensed by the directional coupler 3314increases, the value of the signal V_(SENSE) will decrease. The signalV_(SENSE) is provided to control circuitry 3326. The control circuitry3326 is also provided with a signal P_(SET) which relates to a desiredoutput power level of the power amplifier 3310. The output of controlcircuitry 3326 is provided to the power amplifier 3310 to control theoutput power of the amplifier 3310. The circuit shown in FIG. 33functions to maintain the output power of the power amplifier 3310 at adesired level by sensing the actual output power and adjusting the gainof the power amplifier 3310 accordingly. The gain of the power amplifier3310 is controlled based on the generated signal V_(SENSE) and the valueof P_(SET). The circuit illustrated in FIG. 33 (as well as the circuitsdescribed below) are designed to approximate logarithmic amplifiers.

[0105]FIGS. 34 and 35 illustrate examples of block diagrams of circuitryfor sensing and controlling the output power of an RF power amplifier.FIG. 34 shows a power amplifier 3410 and an antenna 3412 coupled to theoutput of the power amplifier 3410. A directional coupler 3414 iscoupled to the output of the power amplifier 3410 for sensing the outputpower of the power amplifier 3410. The directional coupler 3414generates a signal V_(COUP) which is proportional to the output power ofthe power amplifier 3410. The signal V_(COUP) is provided to a couplerVGA 3418. An optional capacitor C8 is coupled between the directionalcoupler 3414 and the coupler VGA 3418 to provide a DC block. The couplerVGA 3418 illustrated in FIG. 34 is comprised of a multi-stage amplifier.In the example shown in FIG. 34, the coupler VGA 3418 is comprised ofsix linear variable gain amplifier stages 3430. The output of thecoupler VGA 3418 is rectified by a sense circuit 3420. The filtered andrectified signal is provided to a first input to of op amp 3422. Afixed-amplitude DC reference voltage (V_(REF)) is provided to a secondinput of the op amp 3422. The value of V_(SET) is the voltage that isdesired at the output of the sense circuit 3420. The op amp 3422generates a gain control signal based on the inputs to the op amp. Thegain control signal is coupled to each amplifier stage 3430 of thecoupler VGA 3418 and controls the gain of each stage 3430. The gaincontrol signal is also coupled to a conditioning circuit 3424 whichgenerates a DC signal V_(SENSE) which is provided to control circuitry3426. The conditioning circuit 3424 conditions the gain control signalto compensate for the non-linearity of the VGA 3418. The conditioningcircuit 3424 shown in FIG. 34 is comprised of an amplifier 3432 and a DCvoltage source that provides an input voltage V_(REF).

[0106] In the scheme illustrated in FIG. 34,

V _(SENSE) =A _(V) ·V _(REF)  (1),

[0107] where A_(V) is the gain of each stage 3430 of the coupler VGA3418 and is also a function of the gain control signal. Also,

V _(SET) =A _(V) ^(n) ·V _(COUP)(Peak−Peak)  (2),

[0108] where n is the number of stages 3430 of the coupler VGA 3418.Note that equation (2) depends on the implementation used (i.e., whetherthe sense circuit 3420 is comprised of a level detector, an RMSdetector, etc.). Solving for V_(SENSE) gives $\begin{matrix}{V_{S\quad E\quad N\quad S\quad E} = {{A_{V} \cdot V_{R\quad E\quad F}} = {V_{R\quad E\quad F} \cdot {\sqrt[n]{\frac{V_{S\quad E\quad T}}{V_{C\quad O\quad U\quad P}\left( {{P\quad e\quad a\quad k} - {P\quad e\quad a\quad k}} \right)}}.}}}} & (3)\end{matrix}$

[0109]FIG. 36 is a plot of V_(SENSE) versus P_(ANTENNA) (the outputpower of the power amplifier 3410), with six stages 3430. As shown, theV_(SENSE) curve is fairly linear on a log scale. As the number ofamplifier stages 3430 increases, the curve will become more linear.Therefore, for any specific application, the number of stages 3430 usedis determined not only by the gain required, but also by the desiredlinearity (i.e., by the acceptable error level). In one example of acoupler VGA having six stages (e.g., the circuit in FIG. 34) where theamplifier is calibrated at minimum and maximum power levels, the errorfound is plotted in FIG. 37. As shown, the maximum error isapproximately 0.6 dbm. In the example of the specification for GSMdevices, the acceptable error level is ±2 dbm. Therefore, in thisexample, a 0.6 dbm error would be acceptable. If a smaller maximum erroris required, more stages 3430 can be added to the coupler VGA 3418.Similarly, if a larger error can be tolerated, fewer stages 3430 can beused in the coupler VGA 3418.

[0110] One advantage of the present invention is that a 2 point powercalibration is possible, as opposed to calibrating a various powerlevels. Another advantage is that an acceptable accuracy is achieved atlower power levels. Another advantage is that the system is stableindependent of ambient temperature variations. This temperaturestability results from the feedback loop. Since the error curve for adevice is known, another advantage of the present invention is that asimple lookup table can be used to reduce the error even further. Thelookup table may be generated based on the transfer function of thecurve illustrated in FIG. 37.

[0111]FIG. 35 is a block diagram of a circuit similar to the circuitshown in FIG. 34. Unlike the circuit shown in FIG. 34, the circuit shownin FIG. 35 uses an AC reference tone and a second sense circuit toprovide the second input to an op amp. By using the reference tone andthe second sense circuit, simpler circuitry can be used for the sensecircuitry (described below). Like FIG. 34, FIG. 35 shows a poweramplifier 3510 coupled to an antenna 3512 and a directional coupler 3514coupled to the output of the power amplifier 3510 for generating asignal V_(COUP). The signal V_(COUP) is provided to a coupler VGA 3518.The output of the coupler VGA 3518 is rectified and filtered by a sensecircuit 3520. The filtered and rectified signal is provided to a firstinput of op amp 3522. An AC reference tone RFI is provided to a variablelimiter 3534 which limits the peak voltage to the value of V_(SET). Theoutput of the variable limiter 3534 is rectified and filtered by thesense circuit 3528 and is provided to a second input of the op amp 3522.The op amp 3522 generates a gain control signal based on the inputs tothe op amp. The gain control signal is coupled to each amplifier stage3530 of the coupler VGA 3518 and controls the gain of each stage 3530.The gain control signal is also coupled to a conditioning circuit 3524which generates a DC signal V_(O).

[0112] The conditioning circuit 3524 includes first and second sensecircuits 3536 and 3538 which each provide an input to an op amp 3540.The output of the op amp 3540 provides a signal V_(O) to the controlcircuitry 3526 for controlling the output of the power amplifier 3510.At one input to the conditioning circuit 3524, the AC reference tone RFIis provided to a limiting amplifier 3542 which is powered by a voltageV_(X) resulting in an AC signal with a known amplitude (V_(X)). Theoutput of the limiting amplifier 3542 is provided to the second sensecircuit 3538 and to an inverter 3544. The inverter 3544 is powered bythe output (V_(O)) of the op amp 3540, resulting in an AC signal with anamplitude of V_(O). The output of the inverter 3544 is coupled to theinput of a VGA 3546. The gain of the VGA 3546, like the gain of the VGAstages 3530, is controlled by the gain control signal from the op amp3522. The output of the VGA 3546 is coupled to the input of the firstsense circuit 3536. The conditioning circuit 3524 compensates for thenon-linearity of the VGA 3518. Note that, like the sense circuits 3520and 3528, the sense circuits 3536 and 3538 may be matched to improve theperformance of the invention. The signal V_(O) has a function similar tothe signal V_(SENSE) in FIG. 34. The control circuitry 3526 uses V_(O)and P_(SET) to set the output power of the power amplifier 3510 to adesired level. In the circuit shown in FIG. 35, $\begin{matrix}{V_{O} = {\frac{V_{X}}{A_{V}}.}} & (4)\end{matrix}$

[0113] Note that while V_(SENSE) (FIG. 34) is proportional to A_(V),V_(O) (FIG. 35) is proportional to 1/A_(V). This is an advantage as thesignal V_(O) increases as the RF power delivered to the load increases.Note that because of the differences between V_(SENSE) and V_(O), thecontrol circuitry 3426 and 3526 have to be designed accordingly.

[0114] Yet another advantage to the present invention is that the sensecircuitry can be implemented with simple peak detectors or RMS detectorsthat match each other, but do not require absolute accuracy. Thetemperature stability of the circuits shown in the Figures can beimproved by matching the sense circuits. At high frequencies, especiallyin CMOS, it is difficult to build sense circuits that are accurate. Inthe circuit shown in FIG. 35, the sense circuits 3520 and 3528 arematched to enhance temperature stability. In addition, the output of thesense circuit 3520 is compared to the reference voltage V_(SET) (viasense circuit 3428 ). The reference tone RFI may be provided from anexisting signal in the device. For example, RFI could come from thetransmit signal of the power amplifier prior to final stageamplification.

[0115]FIGS. 34 and 35 provide two examples of suitable conditioningcircuits. Many types of conditioning circuits could be used within thespirit and scope of the present invention. For example, in the casewhere the coupler VGA is simply a linear variable gain amplifier, thenthe conditioning circuit may be comprised of a linear device (e.g., awire or a simple gain circuit). In another example, where the couplerVGA has a non-linear function (i.e., the gain is a non-linear functionof V_(coup)) the conditioning circuit may be complicated (e.g., theconditioning circuit 3524 shown in FIG. 35). Therefore, it is evidentthat there are a variety of ways that a conditioning circuit could bedesigned.

[0116] In one embodiment of the present invention, the power amplifier(3310, 3410, 3510 ) and the coupler VGA (3318, 3418, 3518) are formed ona single integrated circuit.

[0117] The power amplifier and coupler VGA may also be packaged usingthe packaging techniques described above (see FIGS. 25-26). For example,a “flip chip” integrated circuit may be mounted to a multi-layer ceramicchip carrier. In one example, all of the components shown in FIGS. 33,34, or 35 except the directional coupler are formed on an integratedcircuit with the directional coupler formed separately, such as on aceramic chip carrier.

[0118] In the preceding detailed description, the invention is describedwith reference to specific exemplary embodiments thereof. Variousmodifications and changes may be made thereto without departing from thebroader spirit and scope of the invention as set forth in the claims.The specification and drawings are, accordingly, to be regarded in anillustrative rather than a restrictive sense.

What is claimed is:
 1. A circuit for generating a control signal forcontrolling the output power of a power amplifier of a wireless deviceby comprising: a power detector coupled to the output of the poweramplifier for generating a detector output signal; and a variable gainamplifier coupled to the power detector for amplifying the detectoroutput signal to a desired level, wherein the value of the controlsignal generated by the circuit is a function of the gain of thevariable gain amplifier.
 2. The circuit of claim 1, further comprising aconditioning circuit coupled to the variable gain amplifier forconditioning an output of the variable gain amplifier, wherein an outputsignal of the conditioning circuit is the control signal.
 3. The circuitof claim 2, further comprising: a sense circuit coupled to an output ofthe variable gain amplifier; and an op amp having an output and firstand second inputs, wherein the first input is coupled to an output ofthe sense circuit and the second input is coupled to a reference signalfor creating a gain control signal at the output of the op amp.
 4. Thecircuit of claim 3, wherein the reference signal is a DC signal.
 5. Thecircuit of claim 3, wherein the gain control signal is used to controlthe gain of the variable gain amplifier.
 6. The circuit of claim 3,wherein the sense circuit is comprised of a first peak detector.
 7. Thecircuit of claim 6, further comprising a second peak detector coupledbetween the reference signal and the second input of the op amp.
 8. Thecircuit of claim 7, wherein the first and second peak detectors arematched.
 9. The circuit of claim 3, wherein the reference signal is anRF signal.
 10. The circuit of claim 1, wherein the power detector iscomprised of a directional coupler.
 11. The circuit of claim 1, whereinthe variable gain amplifier is comprised of a multi-stage variable gainamplifier.
 12. The circuit of claim 11, wherein the variable gainamplifier is comprised of six stages.
 13. The circuit of claim 11,wherein each stage of the variable gain amplifier is controlled by thegain control signal.
 14. The circuit of claim 1, wherein the wirelessdevice is a cellular telephone.
 15. The circuit of claim 1, wherein thepower amplifier and the variable gain amplifier reside together on anintegrated circuit.
 16. The circuit of claim 15, wherein the integratedcircuit is a flip chip semiconductor.
 17. The circuit of claim 16,further comprising a ceramic chip carrier for carrying the flip chipsemiconductor, wherein one or more components of the circuit reside onthe ceramic chip carrier.
 18. The circuit of claim 1, wherein the poweramplifier further comprises: a first switching device connected betweena first supply voltage and a first output node; a second switchingdevice connected between a second supply voltage and a second outputnode; and an inductance coupled between the first and second outputnodes.
 19. A power detector for an RF power amplifier comprising: adirectional coupler for detecting RF power; a variable gain amplifierhaving an input and an output, wherein the input is coupled to thedirectional coupler; a sense circuit coupled to the output of thevariable gain amplifier for rectifying the output of the variable gainamplifier; an op amp for generating a gain control signal based on therectified output of the sense circuit and a reference signal, whereinthe gain control signal is provided to the variable gain amplifier tocontrol the gain of the variable gain amplifier; and conditioningcircuitry for conditioning the gain control signal for use incontrolling the output power of the RF power amplifier.
 20. The powerdetector of claim 19, wherein the RF power amplifier is a poweramplifier for a wireless communication device.
 21. The power detector ofclaim 19, wherein the variable gain amplifier is comprised of aplurality of variable gain amplifiers.
 22. The power detector of claim21, wherein the gain of each of the plurality of variable gainamplifiers is determined by the gain control signal.
 23. The powerdetector of claim 19, wherein the conditioning circuitry compensates forthe non-linearity of the variable gain amplifier.
 24. The power detectorof claim 19, further comprising a second sense circuit coupled betweenthe reference signal and the op amp.
 25. The power detector of claim 24,wherein the level detector and the second sense circuit are matched. 26.The power detector of claim 24, further comprising a variable limitercoupled between the second sense circuit and the reference signal forlimiting the value of the reference signal to a predetermined value. 27.A method of controlling the output power of an RF power amplifiercomprising the steps of: detecting the output power of the RF poweramplifier and generating a first signal from the detected output power;amplifying the first signal using a variable gain amplifier; generatinga gain control signal from the output of the amplifier and from areference signal; using the gain control signal to set the gain of theamplifier; generating a second signal by conditioning the gain controlsignal; and using the second signal to control the output power of theRF power amplifier.
 28. The method of claim 27, wherein the amplifier isprovided by a multi-stage variable linear amplifier.
 29. The method ofclaim 27, wherein the output power is detected using a directionalcoupler.
 30. The method of claim 27, wherein the gain control signal isgenerated by comparing the amplified first signal to a reference signal.31. The circuit of claim 1, wherein a gain control signal controls thegain of the variable gain amplifier, the circuit further comprising aconditioning circuit for generating the control signal, wherein thecontrol signal is related to the reciprocal of the gain control signal.32. The circuit of claim 31, wherein the conditioning circuit furthercomprises: a variable gain amplifier stage controlled by the gaincontrol signal; and sense circuitry combined with a feedback loop,wherein the conditioning circuitry forces an input reference tone levelto be equal to the control signal multiplied by the gain of the variablegain amplifier stage.
 33. An integrated circuit for use with an externaldirectional coupler comprising: an RF power amplifier formed on theintegrated circuit, wherein the integrated circuit is configured suchthat an external directional coupler can be used to generate a detectorsignal based on the output power of the RF power amplifier; and a powerdetector formed on the same integrated circuit to generate an outputsignal based on the detector signal.
 34. The method of claim 33, furthercomprising a ceramic chip carrier for carrying the integrated circuit.35. The method of claim 34, wherein the directional coupler is formed onthe ceramic chip carrier.
 36. The method of claim 33, wherein theintegrated circuit is used for a wireless communication device.